1. Field of the Invention
This invention relates to the radiofrequency reception field in general. More precisely, the invention relates to a method of sampling an analog radiofrequency signal and a device using this method. One particular example application of this invention is the mobile radiocommunication field that typically uses several communication channels within an allocated frequency band, for example radiocommunication systems according to the GSM (Global System Mobile) standard that use a 25 MHz frequency band centered on the 947.5 MHz frequency.
This invention more particularly relates to reception of information which, in such systems, needs be done by extracting only the frequencies of the channel considered to be received, called the useful channel, from the entire frequency band.
The useful channel is recovered by using filtering techniques at the receiver end to eliminate all frequencies located outside the useful channel frequency band, and techniques for converting analog RF signals into digital data streams for digital signal processing.
2. Discussion of the Related Art
In a context of reducing costs of reception systems, there are at least two strategic points; firstly the increase in the degree of integration in order to make a maximum number of elements on silicon to limit the number of external elements, and secondly the possibility of making the entire reception system using CMOS technology with minimum options. Since the CMOS technology is typically the technology used for digital processing, using the same technology to perform analog functions in the reception system means that a single circuit can be used for all digital and analog functions, thus reducing costs.
The state of the art proposes radiofrequency reception systems with frequency conversion with a low intermediate frequency (called “close to zero”) with a so-called “low-IF” architecture, and direct conversion systems with zero intermediate frequency, called “zero-IF”, consisting of transposing the radio signal directly into a base band.
However, when it is required to respect the constraints mentioned above in order to reduce manufacturing costs, each of these architectures, namely the “low-IF” or “zero-IF” introduces distinct problems.
FIG. 1 shows a classical “low-IF” architecture for a radiofrequency signal reception device. This architecture results in interfacing analog radio modules with a digital processing system operating at a relatively low intermediate frequency, close to zero. Information is then processed at this intermediate frequency, so that low pass filters can be used for processing the signal, since pass-band filters are hard to make on a circuit.
More precisely with reference to FIG. 1, the radiofrequency signals received from an antenna 1 pass through a filter 2, that has the function of rejecting frequencies outside the reception band of the system considered. The filter 2 is followed by a low noise amplifier (LNA) 3 to enhance the signal. The output from the amplifier 3 is sent on two channels, namely the I and Q channels respectively each including a mixer 4, 5 for converting the frequency to a low intermediate frequency. To achieve this, the mixer 4 receives a frequency corresponding to the sum of the intermediate frequency and the signal carrier frequency on its second input connected to a local oscillator 6, in the first channel. The effect is to bring the signal to the intermediate frequency. The frequency of the local oscillator is chosen to obtain a low non-zero intermediate frequency. In the second channel, the signal is multiplied at the mixer 5 by the same frequency, and is then phase shifted by 90 [deg.] such that the corresponding outputs of the mixers 4, 5 are in phase and in phase quadrature.
This signal duplication between the I and the Q channels avoids the loss of information that would otherwise occur during aliasing of the image on the signal when the signal is brought to the intermediate frequency and thus eliminates the need for the presence of a band cutoff filter in high frequencies to eliminate the image frequency which is difficult to achieve.
Each mixer 4, 5 is followed by a low pass filter (LPF) 7, 8 respectively to eliminate channels adjacent to the channel that is to be received and parasite frequencies due to aliasing. When a signal being sampled at a sampling frequency Fech includes frequency components with a frequency of more than Fech/2, these frequency components are folded thus possibly submerging the useful signal (phenomenon called “aliasing” in the literature).
Therefore, to avoid the aliasing phenomenon, the passband of the signal to be processed has to be limited to half of the sampling frequency, so as to respect the sampling theorem.
The signal can thus be sampled in 9, 10 without worrying about aliasing and the signal can then be digitized, for example using sigma-delta type analog-digital converters 11, 12 (ADC).
One disadvantage of this architecture is in the production of low-pass filters. Active filtering is used to avoid signal attenuation. The production of amplifiers necessary to make these filters is complex and requires the use of expensive technological options such as double oxide or high density capacitances. Thus in integrated circuits, it is usually not possible to integrate such a filter on the same circuit as the circuit containing the sampling circuit and the digital circuitry, without increasing manufacturing costs.
Also, due to this complexity, technology migrations providing the means of tending to a greater integration density of circuits will be difficult to apply for such an architecture since the size of filters cannot be reduced in the same way as transistors.
Direct conversion architectures, called zero-IF architectures, can limit the disadvantages related to the image frequencies phenomenon. Although zero-IF architectures introduce the same image frequency and IQ channels problem, the difference is that the image frequency is the negative part of the signal that folds onto the positive part. Therefore the power of the image frequency is limited, unlike low-IF in which the image is one of the adjacent channels with a much higher power than the signal.
In the direct conversion system, a frequency corresponding to the central frequency of the reception channel considered, namely the carrier frequency output by the local oscillator, is received on the second input of each of the two mixers 4, 5. Therefore the local oscillator outputs the same frequency (approximately) as the frequency to be received. Therefore mixer outputs 4 and 5 output the signal received in baseband directly provided that the frequency of the local oscillator is chosen such that the intermediate direct conversion frequency is zero.
However, noise at 1/f is a major problem for these architectures, in particular, when the integrated RF circuits of the reception system are made using CMOS technology. The low frequency noise, called noise at l/f, that characterises these transistors is superposed on the converted signal, such that any signal that is too close to zero is submerged. Therefore the use of CMOS technology in the reception system means that the signal cannot be brought to base band before it has been digitized. Therefore the zero-IF architecture cannot be used.
Therefore, a new architecture has been proposed by the Texas Instruments Company for a Bluetooth type receiver based on a new radiofrequency sampling technique, to eliminate active filtering in FIG. 1. This technique is presented in an article entitled “Direct RF Sampling Mixer With Recursive Filtering In Charge Domain” by Khurram Muhammad and Robert Staszewski, presented at the ISCAS 2004 conference held in May 2004. The reader could also usefully refer to documents in US patents 2003/0083852 and 2003/0083033 for further information about the sampling and filtering techniques described below. A multifrequency reception device based on this sampling technique is shown diagrammatically in FIG. 2. Elements in common with the elements in FIG. 1 are marked with the same references. The principle is to transfer sampling 9, 10 in each channel I and Q into the reception system, as close as possible to the antenna. However, the difficulty to be taken into account lies in the aliasing phenomenon due to sampling.
Therefore a filter, called the anti-alias filter, is necessary to take account of this phenomenon. Advantageously, in the solution proposed by the company mentioned above, simple passive filters with switched capacitances may be used, thus replacing active filtering that is difficult to use in traditional architectures described in FIG. 1. Therefore, the function of this type of filter with references 13 and 14 in FIG. 2 for each I and Q channel respectively, is to filter all alias frequencies of the signal before they submerge the signal during sampling. Once the signal has been sampled, it still needs to be filtered to eliminate channels adjacent to the channel to be received, this operation being done by blocks 15 and 16, and it still needs to be undersampled such that the frequency of the analog-digital converter is acceptable.
More precisely, considering an example according to a time representation, consider a signal with frequency fc and a signal with frequency f=fc+n·fs, where fs is the sampling frequency, it can be observed that the two signals have exactly the same value at an interval having a value Ts (sampling period). Thus, when the signal is sampled, the acquired samples cannot be used to find the original signal since solutions with the same frequency are superposed, and in particular the frequency fc cannot be distinguished from frequency fc+fech, frequency fc+2fech, etc. In other words, all frequencies are folded on themselves at the sampling frequency.
However, this aliasing phenomenon has the advantage of reducing the frequency of the signal to be digitized. The highest usable frequency is equal to half of the sampling frequency. Thus, it is impossible to have a useful signal with a frequency more than half the sampling frequency. Sampling of the signal introduces the same spectrum offset by k times fs, in other words at fs, 2fs, etc., in addition to the signal spectrum.
Therefore, the constraint is to provide an anti-alias filter with the role of destroying all parasite frequencies at fc+fs or fc−fs, fc+2s or fc−2fs, fc+3fs or fc−3fs, etc., prior to sampling.
Therefore, the anti-alias filter must be centered on the carrier with its zeros spaced at the sampling frequency. In the solution recommended by Texas Instruments, the anti-alias filter is centered on the zero frequency and has a transfer function A as shown in FIG. 3, describing the cardinal sine function. Therefore it is essential to bring the signal into the baseband before filtering it, so as to use the entire power of the anti-alias filter.
The anti-alias filter shown in FIG. 3 passes through zero, in other words there is no signal attenuation, and the filter includes zeros at every multiple of fs, such that frequencies located at these precise locations of the filter spectrum are cut off. These frequencies are the alias frequencies that are to be eliminated.
The anti-alias filter transfer function is explained as described with reference to FIG. 4, which shows a sampler mixer circuit performing the filtering and sampling functions of blocks 13, 9 and 15, and 14, 10 and 16 respectively as shown in FIG. 2. Each I and Q channel includes such a circuit. The anti-alias filter function is actually included into the sampling operation. Thus, the filter operation filters the received analog signal such that the signal sampled at a given sampling frequency represents a filtered version of the analog signal from which parasite frequencies have been removed.
This is done by firstly converting the RF analog signal into current iRF using the LNA amplifier that is designed to output in current. The current is then sent to either the positive side VIF+ or the negative side V1F− of the circuit, taking account of the pseudo-differential structure that is represented. An LO+, LO− mixer on each side is present to bring the useful signal to zero, multiplying it by the carrier frequency. After the mixer, the current is then integrated into a sampling capacitor reset to zero for each sample. More precisely, a first signal sample is obtained bypassing the current iRF in the first capacitor CR during the sampling period chosen from 0 to Ts. This is done by making the MOS switch SA conducting from 0 to Ts. This is equivalent to integrating the signal over the sampling period after having multiplied it by the carrier frequency. A second signal sample is then obtained by integrating the current in a second capacitor CR during the period from Ts to 2Ts, with the MOS switch SAZ being made conducting from Ts to 2Ts and so on. By integrating the signal over such a time window corresponding to the sampling period, a time convolution is reconstructed. In frequency, this is equivalent to multiplying the signal with a cardinal sine (which is the Fourrier transform of the time window). The transfer function thus produced corresponds to the anti-alias filter in FIG. 3. The filter spectrum made is a cardinal sine centered on zero frequency.
Once the signal has been sampled, it needs to be filtered and possibly undersampled such that the converter frequency is acceptable.
A low pass type filter to eliminate channels adjacent to the useful channel that is to be received is thus made using the capacitor CH in the circuit in FIG. 4 called the memory capacitance, which is charged by the current iRF and that is not intended to be reset to zero. To do the filtering, each acquired sample is averaged with this capacitance CH, one after the other. Thus in a first step, the amplifier outputting in current outputs into the capacitance CR, and in a second step the capacitance CR is averaged with the capacitance CH, which is never reset to zero and therefore keeps the memory of the signal. The spectrum of the filter B is then obtained as shown in FIG. 3 in dashed lines, in which it can be seen that all frequencies around the required signal are filtered. Curve C then represents the combination of these two filter operations: anti-alias filtering and filtering of adjacent channels.
A final filtering operation acting as an anti-alias filter for additional undersampling may also be used. The first sampling is made at a fairly high frequency that is not adapted to the working frequency of the converter. To divide the sampling frequency by N, the last N capacitances CR representative of the last N acquired signal samples, are averaged among themselves. The result is the transfer function as shown in FIG. 5. This filter does not make any attenuation at zero frequency at which the signal is located and includes zeros at all multiples of fs/N, in other words frequencies that could fold on the useful band. In this way, no parasite frequency submerges the signal during sampling.
All operations described above are controlled by clock signals generated by circuits not shown.
All filters presented with reference to FIGS. 2 to 5 are passive filters with switched capacitances, which can therefore be easily made using CMOS technology and can therefore be easily integrated.
As we have already seen, use of CMOS technology introduces the problem of noise at 1/f. Consequently, this noise makes it impossible to bring the signal into the base band. As the distance from the zero frequency decreases, the noise due to MOSs increases. Therefore an intermediate frequency has to be used to offset the useful band from zero, according to the low-IF principle. Thus, in the application chosen by Texas Instruments for a Bluetooth receiver, the signal is firstly brought to a low intermediate frequency, and is then sampled and cleaned by the filters presented above.
However, by choosing an intermediate frequency not equal to zero, the useful signal is completely offset from the spectrum of the filters presented in FIGS. 3 and 5, which are centered on zero. Since these filters are fairly aggressive, as soon as the signal moves away from the peak centered at zero as shown in FIG. 3, it will be significantly attenuated and deformed by filtering. Furthermore, the anti-alias filter will no longer have its zeros at the right location since the intermediate frequency of the signal has been offset. Therefore the signal is significantly modified by filtering and the alias frequencies are cut off less. Therefore, the proposed filters should be offset to match the intermediate frequency. However, filtering would no longer be symmetric. The result would then be imaginary filters in which the I and Q channels would have to be mixed before the filters could be used, which introduces serious problems when it is required to make digital corrections for bad matching of the channels.
Therefore, when making the Bluetooth receiver, a compromise had to be found between a sufficient offset of the useful signal frequency to avoid the noise at 1/f and the filtering power. Such a compromise will be difficult to find, for example for adaptation of such a receiver to the GSM (Global System for Mobile Communications) Standard or the UMTS (Universal Mobile Telecommunication System) standard. In these systems, parasite frequencies and adjacent channels are more powerful, so that more pronounced filtering is necessary. A signal offset from the filter would then be excessively modified. Therefore, the use of this architecture is limited for standards stricter than Bluetooth.
Another problem of this architecture is the image frequency. When the signal is brought directly to zero-IF, the image is exactly symmetric with the signal, which means that the negative part submerges the positive part. The information can then easily be found by combining use of the two channels I and Q. In the case of the “low-IF”, when the signal is brought around an intermediate frequency, the signal is submerged by the adjacent channels. It is then more difficult to eliminate the image frequency and to find the required information using the I and Q channels. According to the standard, as the intermediate frequency increases, the image frequency moves further from the useful band and its power increases. Since matching of the I and Q channel is dependent on the power of the image to be eliminated, matching needs to be very good, particularly when the intermediate frequency is high.